OriginalAmpModuleAnalysis.pdf

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Analyzing the Dynaco Stereo 120 Power Amplifier
The Stereo 120 Power Amplifier came out around 1966. It was the first powerful (60
watts per channel) solid state amplifier in wide production. Each channel was done with
just 6 transistors, so every transistor had to do a lot. This paper analyzes the amplifier’s
design.
A good companion to analysis is a clear drawing of the circuit. The one that Dynaco
provided in their manuals is a bit confusing. I’ve laid out something that I think is easier
to understand in Figure 1 on page 3. Keep a copy of Figure 1 handy as we will refer to it
extensively in the discussion that follows. Note that some of the part numbers were handy
for simulation, but are not the actual types specified in the Bill of Materials.
There’s a lot going on in the amplifier. We’ll break the amplifier into sections, and give
each section its own heading.
Input Circuits
C1, 5 F, is the input blocking capacitor. Assuming the input impedances is around 100K
Ohms, this capacitor makes a high pass filter that is -3 dB at 0.31 Hz, which is way below
the bottom of the audio band. R1 and C2 make a low pass filter that is -3 dB at or below
225 kHz. This can help prevent radio frequency interference on the input.
Main Gain
Q1 and Q2 provide the main gain. Compared to more modern amplifiers, Q1 is
equivalent to the input differential stage, with the input signal applied to Q1’s base and
the feedback signal applied to Q1’s emitter. Q2 performs the VAS function (Voltage
Amplifier Stage). The combination provides large forward gain, such that the addition of
feedback can lower the overall distortion.
C13, 68 pF, limits the main gain stage bandwidth, and is equivalent to Cdom in modern
amplifiers. C15, 27 pF, provides a bit of fast feedback that bypasses the output stage. It
helps amplifier stability some at the cost of a bit of distortion.
Output Stage
The output stage has unity voltage gain at best, but large current gain. Q3 and Q4 are the
output stage drivers. Q5 and Q6 are the output transistors. C4 couples the main gain stage
to the output stage. C3 is a bootstrap capacitor that supplies positive feedback from the
output stage to the main gain stage, increasing the gain of the main gain stage.
C4 is kind of a handy thing. It prevents DC problems in the input stage from wreaking
havoc and destruction on the output stage. Later amplifiers went to direct coupling as
people became more comfortable with transistors.
Page 1 of 5
© Daniel M. Joffe, 2011
C3, the bootstrap capacitor, is also a very inexpensive way to get a lot more forward gain
at a very low price. In today’s amplifiers, C3, R6, and R7 are often replaced by a current
source.
Output Stage Biasing
The output stage biasing was always a bit of a sore point in the Stereo 120. Dynaco
pointed out with some pride that there was no quiescent current in the output stage. At the
time, that was probably a good thing for survivability. Many output stages with quiescent
current were subject to thermal runaway, destroying amplifiers, power supplies, and
speakers. The downside is that the Stereo 120 is prone to crossover distortion.
The output stage biasing loop starts at the bottom of R27, goes through the Q5’s base-
emitter junction, Q3’s base-emitter junction, Q4’s base emitter junction, R28, D3, and
D2. Written as a loop, we have:
Ic5*R27+Vbe5+Vbe3+|Vbe4|+Ic4*R28=Vd3+Vd2
For the moment, assume Ic4 and Ic5 are zero. Then the left side has 3 Vbe’s, and the
right side as two Vd’s (diode voltage drops). This says that the three transistors won’t
have enough voltage to be turned on, at least at room temperature. As you push power
into a load, Q3, Q4, and Q5 warm up. Their Vbe drop at 2 mV/C with increasing
temperature. When they get hot enough, some quiescent current will flow. If they get too
hot, the amplifier might enter thermal runaway. Dynaco tried to prevent that by
specifying the voltage drop across D2 and D3 with 140 mA flowing.
The rise of quiescent current with warmup decreases crossover distortion. This would
certainly make an amp sound more musical. This is perhaps where the high end hi-fi
passion for warmup or break-in got its start. You’d think that modern amps wouldn’t
have this problem, but many amps still have output stage biasing that is far from stable
over temperature.
Page 2 of 5
© Daniel M. Joffe, 2011
;tran 0 2m 1m 1u
R6
1k
10k
0.01
Q3
TIP31
D1
my5p1zener
R7
1.5k
C4
C13
35
68p
D2
1N4004
C15
Q1
2N3904
27p
R5
100k
TIP32c
Q4
C8
0.1
Q6
my2n3772
R8
270
250
R9
4.7k
R10
3.9k
C5
R12
10k
R16
300
R14
100
R29
1k
R28
3.3
Q2
2N5550
R27
0.47
R13
100
50
1N4004
C6
D3
Q5
my2n3772
R15
4.7k
C3
250
300
R3
30k
R11
R17
C14
.ac dec 20 1 1e6
;op
V3
72
C7
3300
L1
10
C1
R1
R2
5
4700
4700
C2
RSPKR
8
R18
4.7
V1
R4
150
150p
SINE(0 0.05 10k)
AC 1
Figure 1 – Schematic of one channel of the Dynaco Stereo 120 Power Amplifier
Page 3 of 5
© Daniel M. Joffe, 2011
Output Stage Protection
D1 and R27 limit positive peaks of load current. At high currents, Q3 might have
Vbe=1.0 volts and Q5 might have Vbe = 1.6 Volts
1
. The positive current limit is then:
5 .1
−
1
−
1 .6
=
5.31
Amps
0.47
R17 limits negative peaks of load current. Assume Q6 is turned on so that its collector is
near ground. The current available to feed Q4, which feeds base current into Q6, is
something less than 70/300=233 mA. Assuming Q6 has β=25 at high collector currents
2
,
the maximum negative load current is 25*0.233=5.83 Amps.
Still, the negative peak current isn’t really well controlled. If Q6 has high β at high
current, then the current limit could approach 15 Amps. The saving grace is that the
current is somewhat limited by the 3300 F output capacitor. After 20 milliseconds with
5 Amps of current, C7 would have completely discharged. Of course, this makes things
harder for Q5 when the input reverses, but the positive peak current limit is much better
defined.
Feedback and Gain
There are two feedback resistors, R9 and R10. R9 is inside C7, and R10 is outside C7.
R10 extends the low frequency cutoff of the amp by making C7 appear larger. Roughly,
the gain of the amplifier from the base of Q1 to the output is given by:
R
9 ||
R
10
Gain
=
1
+
=
15.2
R
4
There’s a slight attenuation at the input owing to a voltage divider between R1, R2, and
R5. That drops the gain by a factor of:
R
5
Input
_
Loss
=
=
0.914
R
5
+
R
1
+
R
2
The overall gain from the input is the product of the two:
Overall
_
Gain
=
15.2
⋅
0.914
=
13.89
The calculated overall gain cross-checks nicely with the rated sensitivity of 1.5 Volts
RMS input for a 60 Watt (8 Ohm) output, corresponding to 21.9 Volts RMS across 8
Ohms. 21.9/13.89=1.57 volts RMS, rather close to the quoted 1.5 Volts RMS.
This large Vbe comes by assuming Ic5=4 Amps, β≈40, so Ib5=0.1 Amps. The base resistance of a
2N3772, according to the model, is 8 Ohms. That adds another 0.8 volts to the intrinstic 0.8V of Vbe at
high currents.
2
The Bill of materials calls for β=60-90 at Ic=1 Amp. Typically, the β droops at high currents, so 25 might
be reasonable around 5 Amps.
1
Page 4 of 5
© Daniel M. Joffe, 2011
It’s interesting to note that the feedback is actually current mode feedback, rather than
voltage mode feedback. This is owing to the use of the single transistor, Q1, rather than a
differential pair.
Cross Conduction in the Output Stage
The user’s manual instructs us to set up high frequency measurements at low levels, then
performing them quickly at high levels. Why is this? The old slow output transistors were
subject to significant cross-conduction at higher frequencies, like a 10 or 20 kHz high
level sine wave input. That made for a current in both Q5 and Q6 that doesn’t appear in
the load, wastes power supply power, and could lead to the failure of the output
transistors. In controlled amounts, it’s called quiescent current. In the undesired cross
conduction case, it can cause the power supply to go into current limiting.
Replacing the 2N3772’s with something more modern, perhaps an MJ15003, greatly
diminishes the amount of cross-conduction at high levels and high frequencies.
Page 5 of 5
© Daniel M. Joffe, 2011

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